A dual-wideband bandpass filter with closely spaced passbands using coupled lines and shunt transmission lines loaded with end-connected coupled lines

Abstract

A fourth-order bandpass filter with dual-wide passbands is proposed in this article. The novelty of the proposed work is the realization of a dual-band bandpass filter with closely spaced passbands, wide passband bandwidth, high rejection between the passbands, using a novel combination of open stub tapped series coupled lines (CLs) and shunt transmission line loaded with end connected CLs. The characteristics of the proposed dual-wideband bandpass filter are investigated by adopting even–odd mode analysis. The transmission zeros generated by the transmission line loaded with an end-connected CL and open stub are utilized to achieve a high skirt rate. A rejection better than 55 dB is achieved with a single transmission zero between the two passbands indicating high isolation and close passbands. The dual-wideband filter is designed and manufactured to operate at 1.33 and 2.32 GHz. The experimented 3-dB fractional bandwidths of the two pass frequency bands are 40.6% and 24.13%. The proposed filter’s tested frequency responses agree well with simulated results. The proposed device can be used in various applications such as telecommunications, satellite communications, radar systems, imaging, and spectroscopy.

Introduction

Lower microwave frequency bands are often allocated for various applications including communication, broadcasting, radar, satellite communication, and more. However, the available spectrum within these bands is finite and limited. As demand for wireless communication services continues to grow, the available spectrum becomes increasingly scarce, leading to congestion. In congested lower microwave frequency bands, the presence of numerous transmitters operating in proximity can lead to interference and noise. To overcome the interference issue for applications with close operating frequencies, there is a need to design dual-band bandpass filters with closely spaced passbands and high rejection between them. Dual-band bandpass filters are characterized by several key parameters which include size, passband frequency, bandwidth, insertion loss, input return loss, and selectivity. There are many different techniques for designing dual-band bandpass filters with coupled lines (CLs), resonators, stubs, and lumped elements [1–30]. Metamaterial structures like open-loop resonators [1], split rings, or complementary split-ring resonators [2, 3] are employed in the design of dual-band filters. These filters occupy large circuit sizes and have poor roll-off rates. A dual-band bandpass filter is designed using W- and V-shaped open transmission lines for WLAN applications and presented in reference [4]. The realized filter has a poor roll-off rate of 17/34 dB/GHz. Another bandpass filter utilizes the unique characteristics of stepped impedance resonators (SIRs) to achieve two frequency bands filtering behavior [5–10]. SIRs are nonconventional resonators that exhibit multiple resonant frequencies due to their stepped impedance structure [5]. This includes the use of SIRs such as a pair of λ/2 bended resonators [6], with folded T-shaped structure and meander line [7], short-circuited SIR [8], shorted and opened stubs loaded by a λ/4 resonator [9] or using stub-loaded SIRs [10, 11]. The disadvantage associated with the dual-band filters based on SIRs is that they have low fractional bandwidths. In reference [12], a microstrip loop resonator loaded with four short-circuited stubs along with direct source-load coupling is demonstrated to develop a 2^nd order dual-band bandpass filter. Multiple resonators loaded with open-stub and coupled feed are employed in reference [13] to realize a dual-band bandpass filter that has a high isolation with wide stopband. However, it occupies large circuit area and has low fractional bandwidth. A high selective dual-band bandpass filter is proposed using a ring resonator loaded with open-circuited stubs in reference [14]. A series transmission line loaded with open-circuited stubs and stepped impedance open/short-circuited stubs is presented in reference [15] to design a compact dual-band bandpass filter operating at Global System for Mobile communication (GSM)and Wi-Fi applications.

Coupled lines are considered a common technique used to design wideband [16–19] and multiband bandpass filters [20–27]. They consist of two or more parallel transmission lines that interact electromagnetically, causing the wave to transfer between them. This coupling effect can be exploited to create resonant structures that exhibit dual-band filtering characteristics. This can be obtained by carefully designing the dimensions and spacing of the CLs, it is possible to achieve the desired passband frequencies and selectivity. Coupled lines offer several advantages for dual-band bandpass filters in terms of wideband, low loss, size compactness, and flexible designs. In reference [20], three transmission lines are coupled to achieve a dual-band filtering response. Coupled lines are also merged with other techniques to improve the filter performance. These include CL along with shorted stubs [21], loading CLs with stepped impedance open stubs [22], the use of two pairs of parallel CLs and two symmetrical step-impedance open-circuited stub [23], the use of series SIRs, parallel open stubs and shunt cascaded CL [24], loading CLs with tri-stepped impedance stubs (SISs) [25], and coupled SIRs [26, 27]. Dual-band filters realized using signal interference technique has advantages of compact size, wide fractional bandwidth, and multiple transmission zeros, etc. Hairpin-line resonators and slot-coupled resonators are employed in reference [28] to develop a multilayer dual-band bandpass filter. A parallel combination of an H-shaped structure using a transmission line with open-circuited stubs and a short-circuited CL are utilized in reference [29] to realize a dual-band bandpass filter with wider passband bandwidths. In reference [30], a flexible dual-band bandpass filter using CLs, SISs, and a transversal filtering section is reported.

This article presents a dual-wideband bandpass filter and the novelty of the proposed work is the realization of a dual-band bandpass filter with closely spaced passbands, wide passband bandwidth, high rejection between the passbands, using a novel combination of open stub tapped series CLs and shunt transmission line loaded with end connected CLs. The transmission line loaded with an end-connected CL produces three transmission zeros. These zeros combine with the wide passband response of the series CLs to form dual passbands with low return loss. An open stub is center-tapped to the above filter configuration to improve the return loss in the passbands and at the same time roll-off rate. The advantages of the presented filter configuration are

• - The two passbands have 3-dB fractional bandwidths of 40.6% and 24.13%.

• - High isolation (rejection better than 55 dB) between the passbands.

• - The passbands are closely placed with a center frequency ratio of 1.74. The 10 dB inter stopband bandwidth is 0.36 GHz.

Dual-band filter analysis and design

The proposed DWBBPF’s transmission line model and its Ansoft Designer circuit simulated S-parameters results are shown in Fig. 1(a) and 1(b), respectively. It consists of a shunt transmission line loaded with an end-connected CL, and a shunt open stub connected in between two series CLs. The circuit simulated magnitude response has five transmission zeros in the stopband and two wide passbands with four poles each. The circuit-simulated upper and lower stopband rejection (LSR) levels are greater than 25 dB with an inter stopband rejection that is higher than 60 dB indicating high passband isolation, and the generated passbands are close. The working principle of the proposed filter configuration can be explained by considering the frequency response of individual elements. The frequency responses of series connected CLs (Z _E1, Z _O1, θ), and transmission line (Z _A, θ) loaded with end-connected CL (Z _E2, Z _O2, θ) are demonstrated in Fig. 2(a) and 2(b), respectively. The frequency response of series connected CLs has a wide passband with three poles and two fixed transmission zeros f _TZ1 and f _TZ5 at 0 and 2f ₀, respectively. The S-parameter response of the transmission line loaded with an end-connected CL contains three transmission zeros f _TZ2, f _TZ3, and f _TZ4. The location of f _TZ2 and f _TZ4 can be varied by changing the impedances of the transmission line loaded end-connected CL. However, f _TZ3 is fixed at f ₀ and can’t be varied. The transmission line loaded end-connected CL is connected to the either side of the series connected CLs, and its response is shown in Fig. 2(c). Three transmission zeros of the transmission line loaded end-connected CL are overlapped with the wide passband response of series connected CLs generating a dual-passband response. However, the return loss and bandwidth are poor. A quarter wavelength shunt open-ended stub which generates a sharp transmission zero at f ₀, as shown in Fig. 2(d) is included in between the series connected CLs to improve the passband input return loss, bandwidth and selectivity, as demonstrated in Fig. 1(b).

Figure 1. Proposed DWBBPF (a) transmission line configuration (b) circuit simulated |S ₁₁| and |S ₂₁|.

Figure 2. Circuit simulated frequency responses of (a) a series connected coupled lines (b) a transmission line loaded with end-connected coupled line (c) a combination of series connected coupled lines and transmission line loaded with end-connected coupled line and (d) an open stub. (Z _E1 = 288 Ω, Z _O1 = 85 Ω, Z _E2 = 240 Ω, Z _O2 = 86 Ω, Z _A = 66 Ω, Z _B = 175 Ω).

The proposed DWBBPF’s circuit simulated frequency responses can be verified theoretically by considering the even–odd mode analysis. The presented symmetrical filter’s even–odd mode schematic models are depicted in Fig. 3(a) and 3(b). The S-parameters for the proposed filter’s even–odd mode input admittances are written in (1) and (2) [24].

(1)\begin{equation}{s_{11}} = \frac{{Y_O^2 - {Y_{OE}}{Y_{OO}}}}{{\left( {{Y_O} + {Y_{OE}}} \right)\left( {{Y_O} + {Y_{OO}}} \right)}}\end{equation}

(2)\begin{equation}{s_{21}} = \frac{{{Y_O}\left( {{Y_{OE}} - {Y_{OO}}} \right)}}{{\left( {{Y_O} + {Y_{OE}}} \right)\left( {{Y_O} + {Y_{OO}}} \right)}}\end{equation}

Figure 3. The schematic of the presented DWBBPF using coupled lines, transmission line loaded with an end connected coupled line and open stub (a) even mode circuit and (b) odd mode circuit.

where

(3)\begin{equation}{Y_{OE}} = \frac{{{Z_{CLE}} + {Z_{TL}}}}{{{Z_{CLE}}{Z_{TL}}}}\end{equation}

(4)\begin{equation}{Y_{OO}} = \frac{{{Z_{CLO}} + {Z_{TL}}}}{{{Z_{CLO}}{Z_{TL}}}}\end{equation}

(5)\begin{equation}{Z_{CLE}} = - j{Z_{C1}}\cot \theta + \frac{{Z_{C2}^2{{\csc }^2}\theta }}{{{Z_{SS}} - j{Z_{C1}}\cot \theta }}\end{equation}

(6)\begin{equation}{Z_{SS}} = - j2{Z_B}\cot \theta \end{equation}

(7)\begin{equation}{Z_{TL}} = Z_{A}\left[ {\frac{{{Z_{SCL}} + j{Z_A}\tan \theta }}{{{Z_A} + j{Z_{SCL}}\tan \theta }}} \right]\end{equation}

(8)\begin{equation}{Z_{SCL}} = \frac{{ - j}}{2}\left( {{Z_{E2}}\cot \theta - {Z_{O2}}\tan \theta } \right)\end{equation}

(9)\begin{equation}{Z_{CLO}} = j\frac{{\left[ {Z_{C2}^2 + \left( {Z_{C2}^2 - Z_{C1}^2} \right){{\cot }^2}\theta } \right]}}{{{Z_{C1}}\cot \theta }}\end{equation}

(10)\begin{equation}{Z_{CL}} = \frac{{{Z_{E1}} + {Z_{O1}}}}{2}\end{equation}

(11)\begin{equation}{Z_{C2}} = \frac{{{Z_{E1}} - {Z_{O1}}}}{2}\end{equation}

The transmission zero frequencies are calculated based on the above equations to verify the presented filter configuration. To find the transmission zeros theoretically, the S ₂₁ in (2) is equated to null, yielding Y _OE = Y _OO. Equating (3) and (4), we get

(12)\begin{equation}\tan \theta \left[ {{Z_{E2}}{{\cot }^2}\theta - \left( {2{Z_A} + {Z_{O2}}} \right)} \right] = 0\end{equation}

From (12), tanθ = 0 for θ = θ _TZ1,5 = 0° or 180°. The corresponding transmission zero frequencies can be calculated from f _TZ1,5 = (θ/90°)f ₀, where f ₀ is the design frequency. Further simplifying the second term of (12), we get

(13)\begin{equation}{\theta _{TZ2,4}} = {\cot ^{ - 1}}\sqrt {\frac{{2{Z_A} + {Z_{O2}}}}{{{Z_{E2}}}}} \end{equation}

From (13), the 2^nd and 4^th transmission zero frequencies are calculated as

(14)\begin{equation}{f_{TZ2,4}} = \frac{{{\theta _{TZ2,4}}}}{{{{90}^ \circ }}}{f_0}\end{equation}

At the design frequency there is an inherent transmission zero f _TZ3 due to the fact that the higher order terms in the denominator of (2) approaches infinity when compared to numerator. Similarly, the transmission pole frequencies of the proposed filter configuration are determined by equalizing S ₁₁ to zero. After simplifying S ₁₁ = 0, an eighth order polynomial equation is obtained as given in (15).

(15)\begin{align}&{P_1}{\cot ^8}\theta + {P_{^2}}{\cot ^7}\theta + {P_3}{\cot ^6}\theta + {P_4}{\cot ^5}\theta + {P_5}{\cot ^4}\theta \nonumber\\ & \quad+ {P_6}{\cot ^3}\theta + {P_7}{\cot ^2}\theta + {P_8}\cot \theta + {P_9} = 0\end{align}

where

(16)\begin{equation}{P_1} = {N_{1E}}{N_{1o}} - Y_0^2\left( {{D_{1E}}{D_{1O}}} \right)\end{equation}

(17)\begin{equation}{P_2} = {N_{1E}}{N_{2O}} + {N_{2E}}{N_{1o}}\end{equation}

(18)\begin{equation}{P_3} = {N_{1E}}{N_{3o}} + {N_{2E}}{N_{2o}} + {N_{3E}}{N_{1o}} - Y_0^2\left( {{D_{1E}}{D_{2o}} + {D_{2E}}{D_{1o}}} \right)\end{equation}

(19)\begin{equation}{P_4} = {N_{1E}}{N_{5o}} + {N_{2E}}{N_{3o}} + {N_{3E}}{N_{2o}} + {N_{5E}}{N_{1o}}\end{equation}

(20)\begin{equation}{P_5} = {N_{1E}}{N_{4o}} + {N_{2E}}{N_{5o}} + {N_{3E}}{N_{3o}} + {N_{4E}}{N_{1o}} + {N_{5E}}{N_{2o}} - Y_0^2{P_{5o}}\end{equation}

(21)\begin{equation}{P_{5o}} = {D_{1E}}{D_{3o}} + {D_{2E}}{D_{2o}} + {D_{3E}}{D_{1o}}\end{equation}

(22)\begin{equation}{P_6} = {N_{2E}}{N_{4o}} + {N_{3E}}{N_{5o}} + {N_{4E}}{N_{2o}} + {N_{5E}}{N_{3o}}\end{equation}

(23)\begin{equation}{P_7} = {N_{3E}}{N_{4o}} + {N_{4E}}{N_{3o}} + {N_{5E}}{N_{5o}} - Y_0^2\left( {{D_{2E}}{D_{3o}} + {D_{3E}}{D_{20}}} \right)\end{equation}

(24)\begin{equation}{P_8} = {N_{4E}}{N_{5o}} + {N_{5E}}{N_{4o}}\end{equation}

(25)\begin{equation}{P_9} = {N_{4E}}{N_{4o}} - Y_0^2\left( {{D_{3E}}{D_{3o}}} \right)\end{equation}

(26)\begin{equation}{N_{1E}} = {Z_{E2}}\left[ {Z_{c2}^2 - {Z_{c1}}\left( {2{Z_B} + {Z_{c1}}} \right)} \right]\end{equation}

(27)\begin{equation}{N_{2E}} = 2{Z_A}\left[ {Z_{c2}^2 - {Z_{c1}}\left( {2{Z_B} + {Z_{C1}}} \right)} \right] - {Z_A}{Z_{E2}}\left( {2{Z_B} + {Z_{c1}}} \right)\end{equation}

(28)\begin{equation}{N_{3E}} = {Z_{E2}}Z_{c2}^2 - {Z_{o2}}\left[ {Z_{C2}^2 - {Z_{C1}}\left( {2{Z_B} + {Z_{C1}}} \right)} \right]\end{equation}

(29)\begin{equation}{N_{4E}} = - {Z_{o2}}Z_{C2}^2\end{equation}

(30)\begin{equation}{N_{5E}} = 2{Z_A}Z_{C2}^2 + \left( {2{Z_B} + {Z_{C1}}} \right)\left( {2Z_A^2 + {Z_A}{Z_{o2}}} \right)\end{equation}

(31)\begin{equation}{D_{1E}} = - {Z_A}{Z_{E2}}\left[ {Z_{C2}^2 - {Z_{C1}}\left( {2{Z_B} + {Z_{C1}}} \right)} \right]\end{equation}

(32)\begin{equation}{D_{2E}} = \left( {2Z_A^2 + {Z_A}{Z_{o2}}} \right)\left[ {Z_{C2}^2 - {Z_{C1}}\left( {2{Z_B} + {Z_{C1}}} \right)} \right] - {Z_A}{Z_{E2}}Z_{C2}^2\end{equation}

(33)\begin{equation}{D_{3E}} = Z_{C2}^2\left( {2Z_A^2 + {Z_A}{Z_{o2}}} \right)\end{equation}

(34)\begin{equation}{N_{1O}} = {Z_{E2}}\left( {Z_{C2}^2 - Z_{C1}^2} \right)\end{equation}

(35)\begin{equation}{N_{2O}} = 2{Z_A}\left( {Z_{C2}^2 - Z_{C1}^2} \right) - {Z_A}{Z_{E2}}{Z_{C1}}\end{equation}

(36)\begin{equation}{N_{3O}} = {Z_{E2}}Z_{C2}^2 - {Z_{O2}}\left( {Z_{C2}^2 - Z_{C1}^2} \right)\end{equation}

(37)\begin{equation}{N_{4O}} = - {Z_{O2}}Z_{C2}^2\end{equation}

(38)\begin{equation}{N_{5O}} = 2{Z_A}Z_{C2}^2 + {Z_{C1}}\left( {2Z_A^2 + {Z_A}{Z_{O2}}} \right)\end{equation}

(39)\begin{equation}{D_{1O}} = {Z_A}{Z_{E2}}\left( {Z_{C2}^2 - Z_{C1}^2} \right)\end{equation}

(40)\begin{equation}{D_{2O}} = {Z_A}{Z_{E2}}Z_{C2}^2 - \left( {2Z_A^2 + {Z_A}{Z_{O2}}} \right)\left( {Z_{C2}^2 - Z_{C1}^2} \right)\end{equation}

(41)\begin{equation}{D_{3O}} = - Z_{C2}^2\left( {2Z_A^2 + {Z_A}{Z_{O2}}} \right)\end{equation}

Equation (15) is solved to obtain electrical length values using a math tool (MATLAB). Four valid electrical lengths (θ _TP1,2,3,4) are obtained, from which the four transmission pole frequencies of the first passband can be calculated using f _TPn = (θ _TPn/90°)f ₀, n = 1, 2, 3, 4. Since the filter response is symmetric, the other four transmission pole frequencies are calculated using f _TP5,6,7,8 = 2f ₀ − f _TP1,2,3,4. Considering Z _E1 = 288 Ω, Z _O1 = 85 Ω, Z _E2 = 240 Ω, Z _O2 = 86 Ω, Z _A = 66 Ω, Z _B = 175 Ω, and f ₀ = 1 GHz, the five transmission zero frequencies and eight transmission pole frequencies are calculated using (12) and (15). The calculated transmission zero frequencies are f _TZ1 = 0 GHz, f _TZ2 = 0.51 GHz, f _TZ3 = 1 GHz, f _TZ4 = 1.49 GHz, f _TZ5 = 2 GHz, and the transmission pole frequencies are f _TP1 = 0.6 GHz, f _TP2 = 0.65 GHz, f _TP3 = 0.75 GHz, f _TP4 = 0.81 GHz, f _TP5 = 1.19 GHz, f _TP6 = 1.25 GHz, f _TP7 = 1.35 GHz, f _TP8 = 1.40 GHz. The calculated frequencies from theory match with the circuit simulations, which verifies the proposed filter design.

The S ₁₁ and S ₂₁ variations for change in coupling factors (K ₁ = [Z _E1 − Z _O1]/ [Z _E1 + Z _O1] & K ₂ = [Z _E2 − Z _O2]/ [Z _E2 + Z _O2]) and the impedances (Z _A & Z _B) of the presented DWBBPF are shown in Fig. 4(a)–(d). In Fig. 4(a), the K ₁ is varied while other parameters are kept constant. As K ₁ increases, the return loss improves in the passbands where the number of poles is increasing. However, the insertion loss in the upper and lower stopbands is decreasing. On the other hand, with the increase of K ₂, the return loss decreases in the passbands and no significant change is observed in the insertion loss throughout the frequency range, as illustrated in Fig. 4(b). With the increase of the impedance Z _A, the input matching in the passbands is getting better and there is no change in the insertion loss, as depicted in Fig. 4(c). However, as K ₂ and Z _A changes, some effects are observed on the 2^nd and 4^th transmission zero frequencies. The return loss and number of poles in the passbands are affected when there is a change in the impedance value Z _B, as shown in Fig. 4(d). The passband bandwidth analysis of the presented DWBBPF using CLs, transmission line loaded with end connected CL and open stub for different parameter variations are shown in Fig. 5(a)–(d). The 3-dB FBWs of the 1^st and 2^nd passbands are increasing when the coupling factor K ₁ and impedance Z _B are increasing. As the coupling factor K ₂ increases, both passbands 3-dB bandwidth decreases. The passband bandwidths increase initially and almost constant as Z _A increases, as shown in Fig. 5(c). Therefore, the passband bandwidths of the proposed DWBBPF can be controlled simultaneously by varying the parameters, but not independently.

Figure 4. The S ₁₁ and S ₂₁ variations for change in (a) K ₁ (K ₂ = 0.47, Z _A = 65 Ω, Z _B = 175 Ω) (b) K ₂ (K ₁ = 0.54, Z _A = 65 Ω, Z _B = 175 Ω) (c) Z _A (K ₁ = 0.54, K ₂ = 0.47, Z _B = 175 Ω) (d) Z _B (K ₁ = 0.54, K ₂ = 0.47, Z _A = 65 Ω).

Figure 5. The passband bandwidth analysis of the DWBBPF using coupled lines, transmission line loaded with end connected coupled line and open stub for various (a) K ₁ (b) K ₂ (c) Z _A and (d) Z _B.

Filter prototype and measurement

The design parameters of the presented DWBBPF are initially estimated from Fig. 5 to satisfy the bandwidth’s requirement. These initial parameter values are fine tuned using a circuit simulator to achieve good passband and stopband characteristics. The finalized impedances of the DWBBPF are Z _E1 = 288 Ω, Z _O1 = 85 Ω, Z _E2 = 240 Ω, Z _O2 = 86 Ω, Z _A = 66 Ω, and Z _B = 175 Ω. The widths and lengths of the quarter wavelength CLs, transmission line loaded with end connected CL and open stub are estimated using a transmission line calculator at the design frequency of 1.85 GHz. The presented filter is analyzed using a full-wave simulation software Ansys HFSS and the layout is shown in Fig. 6(a). The physical dimensions are W ₁ = 0.21, L ₁ = 30.52, W ₂ = 1.02, L ₂ = 29.24, W ₃ = 0.21, L ₃ = 30.93, W ₄ = 0.36, L ₄ = 31.11, W ₅ = 4.6, L ₅ = 7, S ₁ = 0.2, S ₂ = 0.27 (all in mm). Rogers RT 5880 (thickness = 0.79 mm, relative permittivity of 2.2, and tangent loss of 0.0012) substrate is used for manufacturing of the proposed DWBBPF, and the fabricated prototype is depicted in Fig. 6(b). The developed prototype is occupying an area of 0.38λ _g × 0.19λ _g. A vector network analyzer is used to measure the magnitude and group delay responses of the proposed filter. Figure 7(a) & 7(b) shows a comparison between the circuit simulated, full-wave simulated and experimented magnitude of S ₁₁ and S ₂₁. The frequency response contains two passbands separated by an inter-stopband bandwidth (ISB) of 0.44 GHz with a rejection better than 30 dB. The measured 3-dB bandwidth for 1^st passband ranges from 1.06 to 1.60 GHz while the measured 3-dB bandwidth for 2^nd passband ranges 2.04–2.6 GHz. The 3-dB fractional bandwidths of the 1^st and 2^nd passband are 40.6% and 24.13%, respectively. The passbands have measured center frequencies located at f ₁ = 1.33 GHz and f ₂ = 2.32 GHz. The measured insertion loss at f ₁ and f ₂ are 0.62 and 0.8 dB, respectively. Similarly, the tested return loss is 15.3 dB at f ₁ and 18.3 dB at f ₂. The LSR is better than 29 dB and the upper stopband rejection (USR) is better than 25 dB. Considering the 3 and 20 dB attenuation levels, the tested roll-off rates for the lower and upper sides of the first passband (second passband) are 154.5 dB/GHz (340 dB/GHz) and 130.7 dB/GHz (121.4 dB/GHz), respectively. A comparison between the full-wave simulated and tested results for different parameters is shown in Table 1. From Fig. 7 and Table 1, there is an excellent agreement between the simulated and experimented results. Figure 8 shows the group delay responses of the presented DWBBPF. The measured group delay is ranging from 1.71 to 2.97 ns and 1.7 to 2.96 ns for the first and second passbands, respectively. The performance of the proposed DWBBPF using CLs, transmission line loaded with end connected CL and open stub is compared with the recently published similar works in Table 2. Even though some designs are compact when compared to the proposed filter, their fractional bandwidths are less. Overall, the proposed filter has wide fractional bandwidth, high isolation, close passbands, good LSR and USR, and roll-off rate when compared to other reported dual-band filters.

Figure 6. The proposed filter using coupled lines, a transmission line loaded with end connected coupled line and an open stub (a) layout and (b) fabricated prototype.

Figure 7. Magnitude responses of the presented DWBBPF (a) circuit and full-wave simulated (b) full-wave simulated and tested.

Figure 8. The group delays results of the proposed dual-wideband.

Table 1. Comparison between proposed filter simulations and measurements

Table 2. Performance comparison with reported DWBBPF

Conclusion

Dual-band bandpass filters with closely spaced passbands and high rejection between them are in great demand due to the presence of numerous transmitters operating in close proximity at lower microwave frequency regimes. This work proposed a microstrip DWBBPF with a low 10 dB ISB of 0.36 GHz. The filter with closely spaced passbands is realized using a pair of shunt transmission line loaded with end-connected CL, open stub, and series-CLs. The transmission zeros generated near the passbands by the shunt transmission line loaded with an end-connected CL provided a high roll-off rate. The dual-channel response of the proposed has four transmission poles and wide passbands. The tested prototype operates at 1.33 and 2.32 GHz. Even though the passbands are closely placed, it has good isolation with a rejection greater than 55 dB. The developed DWBBPF can be applied in wireless communication systems like Global positioning System and WiMAX applications.